(1) Field of the Invention
The present invention relates to a radio transmission method, radio reception method, radio transmission apparatus and radio reception apparatus and, for example, it relates to a technique suitable for use in a next-generation mobile communication system employing a DFT (Discrete Fourier Transform)-Spread OFDM (Orthogonal Frequency Division Multiplexing) method which is one of radio access modes.
(2) Description of the Related Art
In a next-generation mobile communication system, as the principal characteristics (necessary conditions) needed for radio access modes with respect to an uplink from a mobile terminal (MS : Mobile Station) to a base station (BTS : Base Transceiver Station), there are requirements for a high frequency utilization efficiency and a low PAPR (Peak-to-Average Power Ratio) of a transmission signal. For these requirements to reach satisfaction, the employment of FDMA (Frequency Division Multiple Access) in terms of a single carrier (SC) has come into discussion (for example, see the non-patent document 1 mentioned later). As one of radio access modes which can meet the above-mentioned requirements, special attention has been paid to a DFT-Spread OFDM mode (for example see the non-patent documents 2 and 3 mentioned later).
One feature thereof is that the employment of signal processing in a frequency domain after DFT processing enables a signal component of a single carrier (SC) to be arranged flexibly in the frequency domain.
FIGS. 10(A) and 10(B) each shows an example of arrangement of a signal in a frequency domain. In FIGS. 10(A) and 10(B), RB is an abbreviation of a resource block and signifies, of a system frequency band (system bandwidth), a minimum unit of frequency band each transmission station (for example, MS) uses.
In addition, FIG. 10(A) is an illustration of one example of a case in which RBs are arranged in a state localized (arranged locally) in a system frequency band, where continuous frequencies (subcarriers) are bundled into one RB. On the other hand, FIG. 10(B) is an illustration of one example of a case in which RBs are arranged in a state distributed (arranged distributively) in a system frequency band, where subcarriers with the same RB number #i (i=1, 2, 3, 4, . . . ), arranged discontinuously (at intervals), are bundled, thereby occupying a frequency band corresponding to one RB in FIG. 10(A).
Accordingly, with respect to each of the localized arrangement shown in FIG. 10(A) and the distributed arrangement shown in FIG. 10(B), the occurrence of multi-user interference in the same cell is avoidable in such a manner that each transmission station uses a different RB, which provides a high frequency utilization efficiency. Moreover, in a case in which the frequency scheduling is used simultaneously in the case of the localized arrangement shown in FIG. 10(A), if an RB with a high reception quality is allocated to each transmission station, the throughput of the entire cell is improvable.
Another feature is that, because of a single carrier transmission system, the PAPR is lower than that of a multicarrier transmission system such as OFDM. In addition, the PAPR can be made lower by simultaneous use of a waveform shaping filter (roll-off filter) with a small amount of operations according to windowing processing in a frequency domain.
A detailed description will be given hereinbelow of a DFT-Spread OFDM mode.
FIG. 11 is a functional block diagram taking note of a transmission processing system of a transmission station 100.
This transmission station 100 shown in FIG. 11 is made up of, as a transmission processing system, for example, a turbo encoder 101, a data modulator 102, a DFT (Discrete Fourier Transformer) 103, a subcarrier mapper 104, an IFFT (Inverse Fast Fourier Transformer) 105, a CP (Cyclic Prefix) inserting section 106, a pilot signal generator 107, a DFT 108, a subcarrier mapper 109, an IFFT 110, a CP inserting section 111, a data/pilot signal multiplexer 112, a digital/analog (D/A) converter 113, an RF (Radio Frequency) transmitter 114, and a transmission antenna 115. Moreover, reference numeral 121 designates a reception antenna and numeral 122 depicts a control signal demodulator for demodulating a control signal received through the reception antenna 121 from the reception station 200, each of which is a component of a reception processing system of the transmission station 100.
In the transmission station 100 thus configured, the control signal demodulator 121 demodulates a control signal feedbacked from a reception station 200 and received through the reception antenna 121 so as to extract RB allocation information. The RB allocation information extracted therefrom is supplied to the DFT 103 and further to the subcarrier mapper 104. For example, the RB allocation information includes the numbers of RBs allocated and RB numbers.
On the other hand, a data signal to be transmitted to the reception station 200 is first turbo-encoded (error-correction-encoded) in the turbo encoder 101 and then data-modulated in the data modulator 102 and inputted to the DFT 103.
The DFT 103 carries out DFT processing in units of symbols corresponding to the RB allocation information (the number of RBs allocated) from the control signal demodulator 122 to convert a data signal in a time domain into a signal in a frequency domain. For example, when the number of subcarriers of an RB is taken as NC and the number of RBs allocated is taken as NRB, the DFT processing is conducted in units of NC×NRB symbols.
Furthermore, the subcarrier mapper 104 maps an output signal from the DFT 103 into a subcarrier in a frequency domain under, for example, a localized arrangement [see FIG. 10 (A)] on the basis of the RB allocation information from the control signal demodulator 122, and the IFFT 105 conducts the IFFT processing on the signal in the frequency domain, mapped into the subcarrier in this way, thereby again making a conversion into a signal in a time domain.
For the purpose of principally improving the resistance against the multipath interference, the CP inserting section 106 inserts a cyclic prefix (CP) [equally referred to as a “guard interval (GI)”] into an output signal from the aforesaid IFFT 105 in units of sample (FFT block).
On the other hand, with respect to a pilot signal, the pilot signal generator 107 produces a pilot signal, and the DFT 108 carries out the DFT processing in units of symbols corresponding to one RB.
Moreover, for the purpose of measuring a radio channel quality information (CQI : Channel Quality Indicator) on each RB in the reception station 200, the subcarrier mapper 109 maps an output signal from the DFT 108 into a subcarrier under the distributed arrangement [see FIG. 10(B)] throughout the system frequency band.
The IFFT 110 carries out the IFFT processing on the signal in the frequency domain, thus mapped in terms of the distributed arrangement, so as to again make a conversion into a signal in a time domain, and the CP inserting section 111 inserts a CP into an output signal from this IFFT 110 in units of sample.
Furthermore, the data/pilot signal multiplexer 112 time-multiplexes the data signal from the CP inserting section 106 and the pilot signal from the CP inserting section 111. A time-multiplexed signal (transmission signal) is D/A-converted in the D/A converter 113 and then quadrature-modulated in the RF transmitter 114 to be converted (up-converted) from a baseband signal into a radio frequency signal, and finally transmitted through the transmission antenna 115 to the reception station 200.
FIG. 12 shows an example of arrangement of a data signal and a pilot signal in the above-mentioned transmission signal in the form of a matrix in time and frequency directions.
As mentioned above, in FIG. 12, a domain for a data signal and domains for pilot signals are time-multiplexed, and the domains for the pilot signals are disposed at both end portions of one subframe and the domain for the data signal is disposed therebetween. Moreover, RBs each forms a minimum unit of a frequency band each transmission station uses are arranged locally with respect to the data signal and arranged distributively with respect to the pilot signals. In the example shown in FIG. 12, as indicated by hatching, RB2 is fixedly allocated to the pilot signal of a transmission station A while RB1 and RB2 are allocated to the data signal on the basis of the RB allocation information.
FIG. 13 is a functional block diagram taking note of a reception processing system in the reception station 200.
The reception station 200 shown in FIG. 13 is made up of, as a reception processing system, for example, a reception antenna 201, an RF receiver 202, an analog/digital (A/D) converter 203, a CP deleting section 204, a path searcher 205, a data/pilot signal demultiplexer 206, FFT (Fast Fourier Transformer) 207, 208, a channel estimator 209, a time/frequency interpolator 210, a weighting factor generator 211, a frequency domain equalizer 212, a subcarrier demapper 213, an IDFT (Inverse Discrete Fourier Transformer) 214, a data demodulator 215, a turbo decoder 216, a pilot signal generator 217, a DFT 218, a subcarrier mapper 219, an SIR estimator 220, an RB allocating section 221, a buffer 222, and an effective subcarrier judger 223. Moreover, reference numeral 231 denotes a control signal modulator made to modulate a control signal including the next RB allocation information from the RB allocator 221 and numeral 232 represents a transmission antenna, each of which is a component of a transmission processing system of the reception station 200.
In the reception station 200 thus configured, the RF receiver 202 first converts (down-converts) a radio frequency signal, transmitted from the transmission station 100 and received through the reception antenna 201, into a baseband signal and, after quadrature-demodulated, the A/D converter 203 carries out the A/D conversion thereon.
The digital signal after the A/D conversion is inputted to the CP deleting section 204 and the path searcher 205. The path searcher 205 carries out a correlative operation between a received signal and a replica of a transmission pilot signal (herein after referred to as a “pilot replica”) in a time domain, thereby detecting a reception timing (start point of an effective signal component) of each path.
The CP deleting section 204 deletes the CP from the received signal on the basis of the information on the reception timing detected by the path searcher 205 so as to extract an effective signal component. The extracted effective signal component is inputted to the data/pilot signal demultiplexer 206, thereby demultiplexing a data signal and a pilot signal, time-multiplexed.
In addition, the received pilot signal is inputted to the FFT 208 where it undergoes the FFT processing for the conversion from the signal in the time domain into the signal in the frequency domain and is then inputted to the channel estimator 209. Still additionally, the pilot signal generator 217 generates a transmission pilot replica in the time domain, and this pilot replica undergoes the DFT processing in the DFT 218 for the conversion from the signal in the time domain to a signal in the frequency domain and is then mapped by the subcarrier mapper 219 in terms of the same subcarrier arrangement (distributed arrangement) as that of the transmission station 100.
The channel estimator 209 carries out, with respect to a subcarrier where a pilot signal is arranged in a distributive manner, a correlative operation between a received pilot signal from the FFT 209 and a transmission pilot replica from the subcarrier mapper 219 in a frequency domain so as to estimate a channel distortion of a frequency domain in a radio channel (that is, to obtain a channel estimate).
The SIR estimator 220, as a first purpose, estimates a received SIR to each RB for a data signal on the basis of the channel estimate obtained by the channel estimator 209. As an example of the estimating method, through the use of a channel estimate of a subcarrier where the target pilot signal of the transmission station 100 is disposed for each RB for the data signal, the sum of the square of a real number of the channel estimate expressed by a complex number and the square of an imaginary number thereof is considered to be a desired signal component S and a variance of a plurality of symbols is regarded as an interference signal power I, and the ratio of S and I is set as a received SIR estimate.
As a second purpose, the SIR estimator 220 calculates a noise power estimate to be used in the weighting factor generator 211 which will be mentioned later. Concretely, it is calculated by averaging the respective data signal RB interference powers I, obtained in the process of obtaining the estimate of the received SIR, among the RBs.
The RB allocator 221 allocates an RB, used for the next data signal transmission from the transmission station 100, on the basis of the received SIR estimate of each RB for the data signal. As an example of the allocation method, there is a method of allocating an RB whose received SIR estimate exceeds a specified threshold.
The time/frequency interpolator 210 carries out the interpolation processing (linear interpolation or the like) in a time direction and in a frequency direction on the basis of the channel estimates of portions of subcarriers and FFT blocks in a subframe, obtained by the channel estimator 209, thereby calculating the channel estimates of all the subcarriers and all the FFT blocks in the subframe.
The weighting factor generator 211 generates an MMSE (Minimum Mean Square Error) weight to be used in the frequency domain equalizer 212 which will be mentioned later. For example, with respect to specified subcarriers and FFT blocks, when a channel estimate is taken as H and a noise power estimate is taken as N2, the MMSE weight W is given by the following equation (1) where H* represents a complex conjugate of H.
                    W        =                              H            *                                                                              H                                            2                        +                          N              2                                                          (        1        )            
On the other hand, a received data signal is FFT-processed by the FFT 207 to be converted from a signal in a time domain into a signal in a frequency domain and then frequency-domain equalized by the frequency domain equalizer 212. Concretely, with respect to specified subcarriers and FFT blocks, an operation is made to multiply a received data signal by the aforesaid MMSE weight W corresponding thereto.
The effective subcarrier judger 223 makes a judgment on a position of a subcarrier (effective subcarrier), where an effective data signal is disposed, on the basis of the RB allocation information from the RB allocating section 221, held in the buffer 222.
The subcarrier demapper 213 extracts a signal of an RB, where the target data signal of the transmission station 100 is disposed, from a received signal of each FFT block after the frequency domain equalization by the frequency domain equalizer 212 on the basis of the information on the effective subcarrier judged by the effective subcarrier judger 223.
The IDFT 214 carries out the IDFT processing on the data signal in the frequency domain from the aforesaid subcarrier demapper 213 to make a conversion into a signal in a time domain. The signal in the time domain is data-demodulated by the data demodulator 215 and then turbo-decoded (error-correction-decoded) by the turbo decoder 216, thus providing a data signal restored.
The control signal modulator 231 maps the RB allocation information requested by RB allocator 221 to be used for the transmission of the next data signal from the transmission station 100, obtained by the RB allocator 221, into a control signal and feedbacks it through the transmission antenna 232 to the transmission station 100.
[Non-Patent Document 1] Rui Dinis, et al. “A Multiple Access Scheme for the Uplink of Broadband Wireless Systems”, IEEE Globecom 2004, December, 2004
[Non-Patent Document 2] NTT DoCoMo, “Optimum Roll-off Factor for DFT-Spread OFDM Based SC-FDMA in Uplink” (R1-060318), 3GPP TSG-RAN WG1 Meeting #44, Denver, USA, 13-17 Feb. 2006
[Non-Patent Document 3] Motorola, “Uplink Multiple Access for EUTRA” (R1-050245), 3GPP TSG RAN1 #40 bis Meeting, Beijing, China, Apr. 4-8, 2005
[Non-Patent Document 4] Huawei, “Improved SC-FDMA PAPR reduction by non root-raised cosine spectrum-shaping functions” (R1-051092), 3GPP TSG-RAN WG1 Meeting #42bis, San Diego, USA, 10-14 Oct. 2005
The above description relates to a basic apparatus configuration according to a DFT-Spread OFDM. A description will be given hereinbelow of a case in which waveform shaping filtering is made with the windowing processing in a frequency domain in the transmission station 100 for the purpose of further reducing the PAPR of a transmission signal.
FIG. 14 illustratively shows a processing procedure in a waveform shaping filter (roll-off filter).
First of all, as shown by (1) and (2) in FIG. 14, a signal s(n) is produced by cyclically copying, of an NTX sample signal (signal before the application of a filter) in a frequency domain, each of NTXEXT samples (see oblique line portions) at both ends. In this case, NTXEXT is given using a roll-off rate α, which will be mentioned later, according to the following equation (2).
                              N          TX_EXT                =                  [                                    α              ·                              N                TX                                      2                    ]                                    (        2        )            
Following this, the foregoing signal s (n) is multiplied by a window function in the frequency domain. In a case in which a root raised cosine function shown by (3) in FIG. 14 is used as the window function, a signal k(n) after the application (employment) of a filter (Root cosine roll-off filter) is given by the following equations (3), (4) and (5) [see (4) in FIG. 14].
                              N                      TX            2                          =                              N            TX                    +                                    N              TX_EXT                        ·            2                                              (        3        )                                          f          n                =                                                            1                +                α                                            N                                  TX                  2                                                      ·                          (                              n                +                                  1                  2                                            )                                -                                    1              +              α                        2                                              (        4        )                                                      k            ⁡                          (              n              )                                =                                    s              ⁡                              (                n                )                                      ⁢                                                            1                  2                                ⁢                                  〈                                      1                    -                                          sin                      ⁡                                              [                                                                              π                                                          2                              ⁢                              α                                                                                ⁢                                                      (                                                                                          2                                ×                                                                                                                                        f                                    n                                                                                                                                                                -                              1                                                        )                                                                          ]                                                                              〉                                                                    ⁢                                  ⁢                  (                                    n              =              0                        ,            ⋯            ⁢                                                  ,                                          2                ·                                  N                  TX_EXT                                            -              1                        ,                                          N                                  TX                  2                                            -                              2                ·                                  N                  TX_EXT                                            -              1                        ,            ⋯            ⁢                                                  ,                                          N                                  TX                  2                                            -              1                                )                ⁢                                  ⁢                              k            ⁡                          (              n              )                                =                      s            ⁡                          (              n              )                                      ⁢                                  ⁢                  (                                    n              =                              2                ·                                  N                  TX_EXT                                                      ,            ⋯            ⁢                                                  ,                                          N                                  TX                  2                                            -                              2                ·                                  N                  TX_EXT                                                              )                                    (        5        )            
Therefore, it is known that, with respect to the signal k(n) after the filter application, although the occupied bandwidth becomes larger as the roll-off rate increases, the PAPR becomes smaller.
In this connection, for reducing the PAPR of a transmission signal, a window function other than the Root raised cosine function is also usable. For example, the above-mentioned non-patent document 4 discloses that the PAPR decreases by the employment of a window function which is optimized for each modulation mode and which is not specified by a roll-off rate.
FIG. 15 is a functional block diagram taking note of a transmission processing system of a transmission station 100 to which a waveform shaping filter (roll-off filter) is applied.
The transmission station 100 shown in FIG. 15 differs from the transmission station 100 described above with reference to FIG. 11 in that PSF switching sections 116, 118 and a plurality of waveform shaping filters (pulse shaping filter) 117-1 to 117-NRBall [PSF(1) to PSF(NRBall)] corresponding to the total number NRBall of RBs are provided between a DFT 103 and a subcarrier mapper 104 for a data signal, a waveform shaping filter (PSFp) 119 is provided between a DFT 108 and a subcarrier mapper 109 for a pilot signal, and with respect to each of the data signal and the pilot signal, waveform shaping is conducted between DFT processing and subcarrier mapping processing (unless otherwise specified particularly, the other components marked with the same reference numerals as those used above are the same as or correspond to the components mentioned above.
That is, with respect to a data signal, since the number of output symbols from the DFT 103 varies according to the number NRB of allocated RBs as mentioned above, when the total number of RBs is taken as NRBall, the waveform shaping filters 117-1 to 117-NRBall corresponding to the numbers (1 to NRBall) of allocated RBs are prepared so as to switch the PSF switches 116 and 118 synchronously on the basis of the aforesaid RB allocation information, thereby applying appropriate waveform shaping filters 117-i (i=1 to NRBall).
For example, when the appropriate waveform shaping filters 117-i are applied with respect to a data signal DFT-processed in units of NC×NRB symbols in the DFT 103 as shown by (1) in FIG. 16, a data signal with an occupied bandwidth of NC×NRB×(1+α) made wider (spread) than the occupied bandwidth (NC×NRB) of the effective subcarrier according to the roll-off rate α is obtainable as a filter output as shown by (2) in FIG. 16. Incidentally, the method for the application of the waveform shaping filters 117-i is uniquely determined according to the RB allocation information and is well-known in the reception station 200.
A data signal after the application of the waveform shaping filters 117-i is subcarrier-mapped under a localized arrangement, for example, shown by (3) in FIG. 16 in the subcarrier mapper 104. However, in this example, of NRBall RBs in total, RBs, which are NRB in number, are allocated as effective subcarriers.
On the other hand, with respect to a pilot signal, the number of output symbols from the DFT section 108 is fixed and, hence, the waveform shaping filter 119 is directly applied with respect to an output signal from the DFT 108. For example, when the waveform shaping filter 119 is applied with respect to a signal DFT-processed in units of NC symbols in the DFT section 108 as shown by (1) in FIG. 17, a pilot signal with an occupied bandwidth of NC×(1+α) spread by a quantity corresponding to the roll-off rate α is obtainable as an filter output as shown by (2) in FIG. 17.
In addition, in the subcarrier mapper 109, a pilot signal after the application of the waveform shaping filter 119 is subcarrier-mapped with respect to a system bandwidth [NC×NRBall×(1+α)] under a distributed arrangement, for example, as shown by (3) in FIG. 17.
Furthermore, FIG. 18 is a functional block diagram tasking note of a reception processing system of the reception station 200 to which a waveform shaping filter is applied.
The reception station 200 shown in FIG. 18 differs from the configuration described above with reference to FIG. 13 in that a waveform shaping filter (PSFp) 224 having the same window function (roll-off rate α) as that of the waveform shaping filter 119 on the transmission station 100 side is provided between the DFT 218 for the production of a pilot replica and the subcarrier mapper 219 so as to carry out the same waveform shaping as the waveform shaping by the transmission station 100 side waveform shaping filter 119 at the production of a transmission pilot replica in a frequency domain. Moreover, in this case, the effective subcarrier judger 223 makes a judgment on a position of an effective subcarrier, where a data signal is arranged, by use of RB allocation information while consideration is given to the fact that the application of the waveform shaping filter 117-i in the transmission station 100 widens the occupied bandwidth of a data signal. The components marked with the same reference numerals as those used above are the same as or correspond to the components mentioned above.
FIG. 19 is an illustration of a comparison between a subcarrier arrangement of a data signal shown in FIG. 16 and a subcarrier arrangement of a pilot signal shown in FIG. 17.
In a case in which, as shown in FIG. 19, the waveform shaping filters 117-i and 119 are applied with the same roll-off rate α but in different bandwidths with respect to the data signal and the pilot signal, for example, there is a possibility that a low-quality pilot signal is mapped in the vicinities of both the end portions of an occupied band of the data signal and that, since the channel distortion differs between both the signals, the reception characteristic degrades in the reception station 200.
That is, for example, at the right-side end (higher frequency side) of an occupied band of a data signal, a pilot signal with a low quality (S/N), whose amplitude attenuates due to the waveform shaping filter 119, is mapped with respect to a portion of the effective subcarriers, which can degrade the channel estimation accuracy and deteriorate the reception characteristic. On the other hand, at the left-side end (lower frequency side) of the occupied band of the data signal, with respect to a portion of the subcarriers, a channel distortion due to devices including the waveform shaping filters 117-i and 119 differs between the data signal and the pilot signal, which can cause incorrect channel compensation for the data signal and deteriorate the reception characteristic of the data signal.